Soft transition converter

ABSTRACT

The present invention is a circuit and method for reducing switching and reverse recovery losses in the output rectifiers while creating zero voltage switching conditions for the primary switchers. There are described two output configurations, one employing a soft commutation inductor element a bridge rectifier and a output filter capacitor, the second using a soft commutation inductor element a rectification-filtering bridge composed by tow capacitors and two capacitors. Both secondary circuits can be driven by three primary circuits. A first circuit is a full bridge with phase shift control, and a second circuit is a half bridge topology with an additional bydirectional switch which achieves two goals, on to get soft switching commutation across all the primary switches, the second to create the right waveforms in the secondary suitable with the claims in this invention. The third topology is a phase shifted two transistors forward. The circuits claimed in this invention can provide soft commutation across the primary switching elements and secondary rectifier means, clamping the voltage across the rectifiers to the output voltage eliminating the need for snubbers circuits both in primary and the secondary section.

BACKGROUND OF INVENTION

[0001] 1. Field of the Invention

[0002] This invention relates to DC-to-DC converters, DC-to-AC invertersand AC-to-DC converters. The major characteristic of this powerconversion technique is that primary switching elements switches at zerovoltage and the secondary rectifiers means have negligible reverserecovery losses.

[0003] 2. Description of the Prior Art

[0004] There is a continuing industry demand for increasing powerdensity, which means more power transferred in a given volume. A methodfor increasing the power transfer through the converter is to increasethe switching frequency in order to minimize the size of magnetic andthe capacitors. Using prior art topologies such as forward or flyback,which employ “hard” switching techniques makes high frequency operationless efficient. The switching losses associated with switching elements,which turn on when there is a voltage across them, are proportional withthe switching frequency. An increase in switching frequency leads to anincrease in switching losses and an increase in level of electromagneticinterference (EMI).

[0005] In order to overcome limitations in switching speeds, the priorart has devised a new family of soft transition. The U.S. Pat. Nos.5,132,889 , 5,126,931 , 5,231,563 , 5,434,768 present several methods ofaccomplishing zero voltage switching across the primary switches.

[0006] Another power loss mechanism is due to the reverse recovery inthe output rectifiers. During switching when a negative polarity voltageis applied to a rectifier in conduction the current through therectifier will continue to conduct until all the carriers in therectifier's junctions are depleted. During this period of time thecurrent polarity will reverse, the current flowing form the cathode tothe anode, while the voltage across the diode is still positive from theanode to the cathode. The current flowing in reverse through the diodewill reach a peak value referred in literature as Irrm. Further on,while the rectifiers' junction is depleting the carriers, the rectifierbecomes a high impedance device. The current through the rectifier willdecrease rapidly from lrrm level to zero. During the same time thenegative voltage across the rectifier will build up to high levels.

[0007] During the period of time when there is a negative voltage acrossthe diode and negative current is flowing through it, there will bepower dissipation in the device. This kind of loss is referred in theliterature as reverse recovery losses. The reverse recovery loss isproportional with the reverse recovery current Irrm, the negativevoltage across the rectifier and the frequency.

[0008] The reverse recovery current Irrm, which is a key component inreverse recovery loss, is function of the type of device, thetemperature and the current slope at turn off. The reverse recoverycharacteristics are getting worst for higher voltage rectifiers. As aresult the reverse recovery loss becomes a significant loss mechanismfor higher output voltage applications. The reverse recovery currentlrrm is direct dependent of the current slope at turn off. A soft slopereduces the reverse recovery current and as a consequence reduces thereverse recovery loss. To accomplish a very soft slope current at turnoff an inductive element has to be in series with the rectifier. Theinductor element will prevent a fast current variation dI/dt. Thepresence of an inductive element in series with the rectifier willincrease the negative voltage across the rectifier at turn off. Thereverse voltage across the rectifier can reach very high levels and canexceed the voltage break down of the device, leading to failure.

[0009] RC snubbers or complicated lossless snubers can be added acrossthe rectifier to reduce the reverse recovery loss and the voltage stresson the devices. This leads to complex circuits and which negativelyaffects the efficiency and the reliability. As a result of theselimitations the high voltage converters have to operate at lowerfrequency in order to reduce the power dissipation associated withreverse recovery.

[0010] What is needed is a converter topology which can operate atconstant frequency with zero voltage switching on the primary switchesand soft commutation on the output rectifiers, wherein low current slopethrough the rectifiers at turn off is associated with low negativevoltage across the rectifiers. The lowest voltage across the outputrectifiers in a DC-DC converter is the output voltage. As a result ourgoal is to reduce the negative voltage across the output rectifier tothe level of the output voltage.

BRIEF SUMMARY OF THE INVENTION

[0011] The invention applies to topologies in which the voltage in thesecondary of the transformer has three states. One state wherein thevoltage in the secondary is a positive voltage source, another stagewherein the voltage in the secondary of the transformer is zero with avery low internal impedance and the third stage wherein the voltage inthe secondary of the transformer is a negative voltage source. Toprovide such secondary signal we have identified three topologicalstructures. One is the full bridge phase shifted topology. The secondtopology is a half bridge utilizing and additional bydirectional switch,depicted in FIG. 9A. The third topology is two transistors forward phaseshifted, depicted in FIG. 14A.

[0012] All these topologies operate in a similar manner. An inputvoltage source is applied to the primary of a transformer throughcontrolled switching elements. The primary winding of the transformerhas two terminations. For simplicity we are going to refer to on end ofthe primary winding of the transformer as a dotted end. The secondarywinding of the transformer has also two terminations. When a voltage isapplied to the primary winding of the transformer with the positivepolarity at the dotted end, a voltage will be induced in the secondary.The termination of the secondary winding where the voltage induced has apositive polarity is referred as a dotted end of the secondary winding.

[0013] The input voltage source is applied to the primary winding of thetransformer through two controlled switching elements. Acontrol-switching element is an electronic switch, which can becontrolled by a control signal to exhibit low impedance when theswitching element is turned ON or large impedance when the switchingelement is turned OFF. The input voltage source is applied to theprimary winding through two controlled switching devices, which connectsthe termination of the transformer to the termination of the inputvoltage source selectively. The dotted end of the primary winding of thetransformer can be connected to the positive end of the input voltagesource and the other end of the primary winding of the transformer isconnected to the negative end of the input voltage source. Thisoperation will be further referred to as positive voltage across theprimary winding.

[0014] The controlled switching elements can connect also the dotted endof the primary winding of the transformer to the negative end of theinput voltage source and the other end of the primary winding of thetransformer is connected to the positive end of the input voltagesource. This operation will be further referred as negative voltageacross the primary winding.

[0015] The control switching elements can also short out the primarywinding of the transformer by applying low impedance across the winding.This operation will be further referring to as the dead time.

[0016] The controlled switching elements can be controlled in a such wayto apply sequentially a positive voltage across the primary winding forgiven period of time, referred as positive ON time, short the primarywinding for a period of time, referred as dead time, apply a negativevoltage across the primary winding for a given period of time, referredas negative ON time, equal as duration with the positive ON time. If thesummation of positive ON time, dead time and negative ON time isconstant, the mode of operation is referred as constant frequencyoperation.

[0017] The power converter can also operate in frequency modulationmode, wherein the summation of positive ON time, dead time and negativeON time is not constant. We introduce the term of duty cycle, which isdefined as the ratio between the summation of positive and negative ONtime and the summation of the positive ON time, twice the dead time andnegative ON time. By varying the duty cycle the power transferredthrough the transformer can be controlled. The duty cycle can be variedby varying the duration of the positive and negative ON time, for theconstant frequency operation. For variable frequency operation the dutycycle control can be made by maintaining the negative and positive ONtime constant and varying the dead time, or by varying the positive andnegative ON time and maintain the dead time constant or by varying thepositive and negative ON time and the dead time in the same time.Important is to have the positive ON time equal to the negative ON time.Another important element of this technology is the low bydirectionalimpedance across the secondary winding of the transformer, wherein thesecondary current can flow freely in both directions.

[0018] One key element in this invention is an additional inductorelement in series with the secondary winding, labeled soft commutationinductor. The inductor can be also located in the primary section inseries with the primary winding of the transformer. The soft commutationinductive element can be also split, one section located in the primary,in series with the primary winding and an another section in thesecondary in series with the secondary winding. In the case when thesoft commutation inductor is located in the secondary, there is abydirectional rectification means connected in series with it and thesecondary winding. A bridge of rectifiers can form the bydirectionalrectification means. Across the capacitor element is connected the load.The bridge of rectifiers has a first input terminal a second inputterminal a first output terminal and a second output terminal. The firstrectifier is connected between the first input terminal and the firstoutput terminal with the cathode to the first output terminal, thesecond rectifier is connected between the second input terminal and thefirst output terminal, with the cathode to the first output terminal.The third rectifier is connected between the second input terminal andthe second output terminal with the cathode to the second inputterminal, the fourth rectifier is connected between the first inputterminal and the second output terminal with the cathode to the firstinput terminal. The AC voltage source in series with said inductiveelement is connected between the first input terminal and the secondinput terminal. The output capacitor is in parallel with the load isconnected between the first output terminal and the second outputterminal.

[0019] The bydirectional rectification means can be also constructedusing two rectifiers and two capacitors. The bridge of rectifiers meansand capacitors having a first input terminal a second input terminal afirst output terminal and a second output terminal. The first rectifieris connected between the first input terminal and the first outputterminal with the cathode to the first output terminal, the secondrectifier is connected between the first input terminal and the secondoutput terminal, with the cathode to the first input terminal. The firstcapacitor is connected in between the first output terminal and thesecond input terminal, and the second capacitor is connected between thesecond input terminal and the second output terminal. The said ACvoltage source is in series with the soft commutation inductive elementand connected between the first input terminal and the second inputterminal. The load is connected between the first output terminal to thesecond output terminal.

[0020] During the positive and negative ON time the power is transferredfrom the primary to the secondary via the transformer, the softcommutation inductor, and the bydirectional rectifier means to the load.In the same time energy is stored in the soft commutation inductor.During the dead time, the energy stored in the soft commutation inductoris further transferred to the load. There are two modes of operation.One mode of operation referred as discontinuous conduction mode, theentire energy stored in the soft commutation inductor is transferred tothe load prior the change of the voltage polarity on the transformer.The second mode of operation referred as continuous mode, there is stillenergy left in the soft commutation inductor prior the reversal of thevoltage polarity in the transformer. The discontinuous mode of operationhas the advantage of transferring the energy from the primary to thesecondary unidirectional at each cycle. The continuous mode of operationwill transfer the energy left in the soft commutation diode back to theprimary before the next energy transfer from primary to the secondarystarts.

[0021] A critical conduction mode of operation can be implementedwherein the reversal of the voltage polarity in the transformer isaccomplished just after the entire energy in the soft commutationinductor is transferred to the secondary. This leads to a modulation infrequency, wherein the frequency will increase at light loads, anddecrease at heavy loads. A mix mode of operation can be also implementedwherein some high frequency boundary or low frequency boundaries or bothare set. There are several major advantages of this topology.

[0022] One of the major advantages is the fact that the voltage acrossthe rectifiers is clamped to the output voltage. There is not ringing orspike across the rectifiers which exceed the output voltage. The voltageacross the rectifiers for a given output voltage is the lowesttheoretical possible. In most of the topologies operating over a rangeof input and output voltages the voltage across the rectifiers can beseveral times larger that the output voltage. For a single ended forwardconverter the output voltage is Vr=(Vin_(Max)/Vin_(Min))*V_(O)/D_(Max),wherein Vin_(Max) & Vin_(Min) is the maximum and minimum input voltageand D_(Max) is the maximum duty cycle, and Vo is the output voltage. Foran input voltage range of 2:1 and 50% maximum duty cycle, the reversevoltage across the rectifier is 4*Vo . In conclusion in these topologieswe achieve the lowest voltage across the rectifiers for a given outputvoltage.

[0023] Another major advantage of this topology is the fact that thecurrent slope through the rectifier at turn off is controlled by thesoft commutation inductor. As a result there is a controlled dI/dt. Asoft current through the rectifier at turn OFF reduces considerably thereverse recovery current. The clamped voltage across the outputrectifiers in association with the soft current slope at turn OFF leadsto reduced reverse recovery losses. If the circuits operate incontinuous mode the reverse recovery losses are reduced, and if weoperate in discontinuous conduction mode the reverse recovery losses areactually eliminate. This is due to the fact that the current through therectifiers reaches zero prior the reverse voltage is applied to them.

[0024] The invention can be better visualized by turning to thefollowing drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

[0025]FIG. 1 is a schematic of a converter utilizing the power transfermethodology of the invention.

[0026]FIG. 1B is a timing diagram of the circuit of FIG. 1.

[0027]FIG. 2 is a schematic diagram of an AC-DC converter wherein priorart technique is illustrated.

[0028]FIG. 3A is a schematic diagram of a DC-DC Converter whereinanother prior art technique is illustrated.

[0029]FIG. 3B is a timing diagram of the circuit of FIG. 3A.

[0030]FIG. 4A is a schematic diagram of an AC-DC Converter using anembodiment of this invention.

[0031]FIG. 4B is a timing diagram of the circuit of FIG. 4A.

[0032]FIG. 5A is a schematic diagram of an AC-DC Converter using anotherembodiment of this invention.

[0033]FIG. 5B is a timing diagram of the circuit of FIG. 5A.

[0034]FIG. 6A is a schematic diagram of an AC-DC Converter using anembodiment of this invention depicted in FIG. 4A, operating incontinuous mode.

[0035]FIG. 5B is a timing diagram of the circuit of FIG. 6A.

[0036]FIG. 7A is a schematic diagram of an AC-DC Converter using anotherembodiment of this invention.

[0037]FIG. 7B is a timing diagram of the circuit of FIG. 7A.

[0038]FIG. 8A is a schematic diagram of a DC-AC Converter, utilizing aphase shift bridge topology suitable with the AC-DC converters depictedin several embodiments of the invention.

[0039]FIG. 8B is a timing diagram of the circuit of FIG. 8A.

[0040]FIG. 9A is a schematic diagram of an DC-AC Converter, utilizing ahalf bridge topology employing an additional bydirectional switch,topology suitable with the AC-DC converters depicted in severalembodiments of the invention.

[0041]FIG. 9B is a timing diagram of the circuit of FIG. 9A.

[0042]FIG. 10A is a schematic diagram of an AC-DC Converter depicted inFIG. 4A wherein two of the rectifiers are replaced by synchronousrectifiers.

[0043]FIG. 10B is a timing diagram of the circuit of FIG. 10A.

[0044]FIG. 11A is a schematic diagram of an AC-DC Converter depicted inFIG. 4A wherein all of the rectifiers are replaced by synchronousrectifiers.

[0045]FIG. 11B is a timing diagram of the circuit of FIG. 11A.

[0046]FIG. 12A is a schematic diagram of an AC-DC Converter depicted inFIG. 5A wherein all of the rectifiers are replaced by synchronousrectifiers.

[0047]FIG. 12B is a timing diagram of the circuit of FIG. 12A.

[0048]FIG. 13A is a schematic diagram of an DC-DC Converter using theembodiment presented in FIG. 4A, wherein the soft commutation inductoris transferred in primary section in series with the primary winding.The AC signal across the primary winding connected in series with thesoft commutation inductor is produced by a phase shift full bridgetopology depicted in FIG. 8A.

[0049]FIG. 13B is a schematic diagram of an DC-DC Converter using theembodiment presented in FIG. 4A, wherein the soft commutation inductoris split in two sections, one section transferred in primary section inseries with the primary winding, and the other section in the secondary.The AC signal across the primary winding connected in series with onesection of the soft commutation inductor is produced by a phase shiftfull bridge topology, depicted in FIG. 8A. This structure being anotherembodiment of the invention.

[0050]FIG. 13C is a schematic diagram of an DC-DC Converter using theembodiment presented in FIG. 4A, wherein the soft commutation inductoris transferred in primary section in series with the primary winding andproviding a center tap, connected to a capacitor. The AC signal acrossthe primary winding connected in series with one section of the softcommutation inductor is produced by a phase shift full bridge topology,depicted in FIG. 8A. This structure being another embodiment of theinvention.

[0051]FIG. 13D is a schematic diagram of an DC-DC Converter using theembodiment presented in FIG. 4A, wherein the transformer is implementedby using two identical transformers connected in series. The additionalcircuit formed by an inductor in series with the capacitor is connectedin between the primary ground and the connection between the two primarywindings of the two transformers. The AC signal across the primarywindings is produced by a phase shift full bridge topology, depicted inFIG. 8A. This structure being another embodiment of the invention.

[0052]FIG. 14A is a schematic diagram of an DC-DC Converter using theembodiment presented in FIG. 4A, The AC signal across the primarywinding of the transformer is produced by two phase shifted twotransistor forward topologies. This structure being another embodimentof the invention.

[0053]FIG. 14B is a timing diagram of the circuit of FIG. 14A.

[0054]FIG. 15 presents a magnetic-packaging structure suitable for theimplementation of the embodiments of the invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0055] For the AC-DC and DC-DC converters for high voltage applicationwherein the use of Schottky rectifiers is not possible due to highvoltage across the rectifiers, one of the major obstacles is the reverserecovery loss of the rectifiers. The reverse recovery losses areproportional with the current slope through rectifier at turn OFF, thereverse voltage across the rectifier after the rectifier exhibits highimpedance, the temperature, frequency of operation and the devicecharacteristics. Additional snubber circuits are often used to reducethe voltage stress on the rectifiers during switching. The frequency ofoperation has to be also reduced which leads to poor volumetricefficiency of the converters. In FIG. 2 are presented a prior art anAC-DC converter. An AC source drives the primary of the transformer T6,950. There is inherent leakage inductance in the transformer, which actsas a current source during commutation of the rectifiers 406 and 408. Inthe right side of the rectifier there is another current source formedby 410. During switching transitions there are two unmatched currentsources on both ends of the rectifiers. This leads to high voltagespikes across the rectifiers 406 and 498. Additional RC snubbers arenecessary to maintain the voltage across the rectifiers below thebreakdown voltage. This leads to higher power dissipation and high EMI.

[0056] In FIG. 3A there is presented another prior art topology used forhigh voltage applications wherein an additional Lv. 12, is added in theprimary of the transformer Tr1. Additional clamping diodes D5 and D6 arealso inserted. The reverse recovery losses in the secondary rectifiersare reduced due to controlled current slope during switching. Thelimitation of the current slope is due to Lv. The presence oftwo-unmatched current source such as the Lv and 30 leads also to voltagespikes across the secondary rectifiers 22, 28, 24 and 26. This requiressnubbers circuits across the output rectifiers, which will reduce theefficiency of the converter.

[0057] This invention proposes several circuits wherein there is notneed for snubber circuits across the secondary rectifiers, and thereverse recovery losses are minimized and under certain conditionseliminated. The maximum voltage across the rectifiers in this inventionis clamped to the output voltage. This will allow a very efficientoperation at high frequency in high power and high voltage applicationsleading to an increase of the power density of the converter.

[0058] In FIG. 4A is depicted an AC voltage source Vs, 46. The voltagesignal produced by 46 is presented on time diagram 60 in FIG. 4B. FromTo to T1 the voltage source 46 provides a positive voltage, 154. Theduration of this signal is referred as Ton. The Ton is the differencebetween T1 and To. After T1 the voltage source 46, provides a zeroamplitude voltage signal 156, exhibiting in the same time very lowinternal impedance. During the time interval T1-T2 , the voltage source46 acts as a sort circuit. The current can flow bydirectional through46, without developing a significant voltage drop. After T3, the voltagesource 46, provides a voltage 158 of opposite polarity to 154 and thesame amplitude. The time interval T3 to T4 when 46, provides 158, isequal to the time interval To to T1. The product of the time interval(T1-To) and the voltage amplitude 154 is equal to the product of thetime (T4-T3) and the voltage amplitude of 158. After T4, the voltagesource 46, is equivalent again to a short circuit providing a zeroamplitude voltage. This state will apply for the time interval betweenT4 to T6. The time interval T1 to T3 and the time interval T4 to T6 arepreferred to be equal, though it is not necessary. The embodiment ofthis invention will also apply if T3-T1 is not identical to T6-T4.

[0059] This type of voltage source can be produced by a transformer andseveral controlled switchers as is depicted in FIG. 8A, 9A and 14A. Themethod of control pictured in 8A is well known in the prior art.

[0060] As is depicted in FIG. 4A in addition to the voltage source 46,there is an inductive element Lr, 48 and a bridge of rectifiers, 900,formed by rectifiers 50, 54, 52, and 56. The inductive element Lr isreferred in this application as the soft commutation inductor element.The voltage source 46 is connected in series with inductor 48, andapplied across the rectifier bridge 900 with one termination to thecathode of 56 and the anode of 50. The second termination goes to thecathode of 52 and the anode of 54. A capacitor 58 is connected acrossthe other two termination of the bridge, respectively between the anodesof 56 and 52 and the cathodes of 50 and 54. A load 902 is connectedacross the capacitor 58.

[0061] During To to T1 when 46, produces a positive voltage 154 at theend connected to Lr, the current fill flow through Lr, rectifier 50,rectifier 52 the capacitor 58 and the load 902. It is assumed that thevoltage across 58 exhibits low voltage ripple much lower than thevoltage amplitude produced by 46. With this assumption the currentthrough Lr will build up linearly as pictured in 62 of FIG. 4B, fromzero to a peak value at T1. During this period of time the energyprovided by 46, will be transferred to 902, and a part of energy willremain stored in Lr. During the time interval between T1 to T2 when thevoltage source Vs is equivalent to a short circuit, the current throughLr will continue to flow until the entire energy stored in Lr will betransferred to the load 902. This will occur at T2, when the currentthrough Lr will reach zero. During the interval T2 to T3 there is notenergy transferred to the load. Do1 and Do1′ ceases to conduct after thecurrent through Lr reaches zero. A voltage ringing may occur across therectifiers of bridge 900, as a result of the resonant circuit formed byLr and the parasitic capacitance of the rectifiers, 50, 52, 54 and 56.The energy contained in this resonant circuit is very small, if theparasitic capacitance of the rectifiers is small. A saturable inductorelement is in series with Lr may damp this oscillation if its energycontent is too high. In the critical conduction mode of operationwherein the voltage produced by Vs will change polarity after thecurrent through Lr reaches zero this ringing will not exist. The timeinterval between T2 and T3 can be controlled in such way that thepolarity change of the voltage produced by 46 will occur after theringing across the rectifiers forces the second set the rectifiers,which will conduct in the next cycle, in this case 56 and 54, inconduction. In this mode of operation there will be zero voltageswitching for the rectifiers. The disadvantage of this mode of operationis the frequency modulation for load and line change. The advantage ofthis mode of operation is that we create ideal switching condition forthe rectifiers. A combination of critical conduction through Lr and fixoperating frequency for light load operation it may give the optimumoperating mode. At the moment T3, the voltage source 46, changes itspolarity and the current fill start flowing linearly through Lr in theopposite direction. At the moment T4, when the voltage produced by 46,becomes zero, the current through Lr reaches its peak. Further betweenT4 to T5 the entire energy stored in Lr will be transferred to the load902. This will occur at the moment T5 when the current through Lrreaches zero. In between T5 and T6 we have the same mode of operation asdescribed between T2 and T3.

[0062] The advantage of this mode of energy transfer from the source 46,to the load 902 is the fact that the voltage across all the rectifierscontained in the bridge 900 is clamped to the voltage across Co which isalso the output voltage of the converter and the fact that the currentthrough the rectifiers at turn OFF is zero. The voltage across all therectifiers reverses only after the current reaches zero. This mode ofoperations eliminates reverse recovery losses, which is a significantadvantage for high output voltage application. If the mode of operationis critical conduction through Lr, and the voltage commutation of 46,occurs after the current through Lr reaches zero and delayed by a timeinterval until the voltage across the rectifiers which will conduct atnext cycle to reach zero, or its lowest level. The control of thevoltage reversal of the voltage source 46 can be done through anagogiccircuits which sense the current through Lr and the voltage across therectifiers, or can be done with microcontroller using digital signalprocessing.

[0063] In FIG. 6A there is presented the same configuration as FIG. 4A.The difference is the mode of operation, which is depicted in FIG. 6B.After a positive voltage is applied by 46, and the current is buit upthrough Lr, until the voltage source 46, becomes a short circuit. Thecurrent will continue to flow through Lr transferring its stored energyto the load. When the voltage produced by Vs changes its polarity thereis current present through the Lr. The voltage across Lr is the sumbetween the output voltage 904 and the voltage across Vs, 158. Thecurrent through Lr will decrease at a higher rate and will reach zero atT8. Between T3 and T8 the current will force the rectifier 50 and 52 tocontinue to conduct. The reverse of conduction will occur at T8, whenthe rectifier 56 and 54 will be forced into conduction. In this mode ofoperation there are reverse recovery losses due to the conduction of therectifiers when the reverse voltage is applied to them. The reverserecovery losses are not high due to the fact that the rate of currentchange through rectifiers at turn off is limited by Lr. In addition tothat the reverse voltage across the rectifiers is clamped to the outputvoltage 904. During the time interval T3 and T8 and T6 to T9, the energyis transferred from the Lr to the voltage source. In conclusion duringT7 to T1 and respectively T8 and T4 the energy is transferred from theprimary to the load and to the Lr. During the time interval To to T7 andrespectively T3 and T8, the energy is transferred from Lr to the source.This kind of operation is less efficient that the one presented in FIG.4A. In most of applications this mode of operation referred ascontinuous conduction mode can occur during the transient situations.The current through the Lr is depicted in 162. The voltage across therectifiers is depicted in 164 and 168. The current through 50 and 52 isdepicted in 166. In 170 is depicted the current through 56 and 54.

[0064] It is very important to underline the fact that the key advantageof this invention is the energy transferred from primary to secondary isdone in one direction only, if the operation occurs in discontinuousmode. The voltage source Vs would not exhibit short circuitcharacteristics during T1 to T3 and T4 to T5, the energy will be alsotransferred from the load to the primary. This limits the numbers oftopologies, which can be used in the primary side, capable to providethe right characteristics for Vs.

[0065] In the circuit presented in FIG. 5A and 6A the energy transferredfrom the Vs to the load is function of the voltage across Vs, the outputvoltage 904 and the inductor Lr. The circuit 7A presents a method ofpower transferred wherein the amount of energy transfer for a given Vs,Lr and Vo can be further increased and modulated by the additionalbydirectional switch S2. The additional switch S2 does not have to bebydirectional.

[0066] The bydirectional switch will maintain the symmetry of the energytransfer for the positive cycle 154 and negative cycle 158. Aunidirectional switch will modulate the power only for positive ornegative cycle. The timing diagram on FIG. 7B is referring to abydirectional switch. The control signal for the switch S2, 336, issynchronized with the voltage source Vs. The switch is turned ON when apositive polarity voltage 154, and a negative polarity voltage 158 isramping up from Vs. During the time interval from To to T10, a positivevoltage 154 is provided by Vs, and the switch 336 is ON. As a result thecurrent will flow through Lr with a slope determined by the amplitude of154 and the value of Lr. The value of the output voltage 904 does notplay any role. The current slope in between To and T10 is high and moreenergy is stored to Lr during this interval. When the switch S2 isturned OFF at the moment T10 the current which flows through Lr willturn ON the rectifiers 50 and 52, transferring the energy to the load902. The current continues to build up through Lr at a smaller rateimposed by the difference between the amplitude of 154 and the outputvoltage. At the moment T1 the current through Lr reaches its peak. FromT1 to T2 all the energy contained in Lr will be transferred to theoutput if the unit operates in discontinuous mode, as is depicted inFIG. 7B. The unit can also operate in continuous mode, wherein thecurrent will not reach zero before the voltage Vs will reverse itspolarity. When the voltage Vs changes its polarity at T3, the switch S2will be also turned on and the current will build up through Lr with thesame slope as between To to T10, but of opposite polarity. The majoradvantage of this embodiment over the embodiment presented in FIG. 4Aand 6A, is the fact that the energy transfer from Vs to Lr isindependent of Vo during the To to T10, respectively T3 to T12 period.The energy transferred during a given time is higher without increasingthe amplitude of 154. The output power can be also modulated not only bythe ratio of the ON time which is the summation of the (T1-To) and(T4-T3) and the period of the signal depicted in 160. The output powercan be further modulated by the ON time of S2.

[0067]FIG. 10A is an extension of the embodiment depicted in FIG. 4Awherein the rectifiers 56 and 52 are replaced by the controlledsynchronized rectifiers M10, 802, and M11, 804. These synchronousrectifiers are controlled by the Vc2 and Vc1 depicted in 700 and 702.The use of synchronous rectifiers may have the advantage of lowervoltage drop, which translates in a higher efficiency. Between To to T1when a positive voltage 154 is applied to Lr, the current will flowthrough Do1 and M11. The control signal 908 turns ON the M11 at To. Thecontrol signal 908 has to keep M11, turned ON until T2. The controlsignal Vc1 can be maintain high even after T2 without impacting the modeof operation. This is a major advantage for using synchronous rectifiersbecause the timing at turn OFF is not very critical. Important is toturn OFF, Vc1 prior the T3.

[0068] Another embodiment of this invention is presented in FIG. 11A.The bridge 900 is formed by the synchronous rectifiers controlled bycontrolled signals Vc1 and Vc2. The timing diagram for Vc1 and Vc2 aredepicted in FIG. 11B in 704 and 706. The mode of operation for theconverter depicted in FIG. 11A is the same as the one depicted in FIG.4A for discontinuous and critical conduction mode and the one depictedon 6A for the continuous conduction mode.

[0069] Another embodiment of the invention is presented in FIG. 5A. Thecircuit formed by the voltage source Vs and the Lr in series isconnected to a bridge formed by two rectifiers 134 and 136 and twocapacitors 138 and 140. The output voltage 904 is the voltage across thetwo capacitors 138 and 140. The output voltage is further applied to aload. In this there are used only two rectifiers. Between To and T1 thevoltage source Vs provides a positive polarity voltage 154 at the endconnected to Lr. The current will flow through Lr, Do1 and Co1. At T1the current through Lr reaches its peak. Between T1 to T3 the source 46becomes a short circuit 156. The current continues to flow through Lruntil reaches zero at T2. At that time all the energy stored in Lr istransferred to the Co1 and Load, via Vo. At T3 the voltage polarityproduced by 46 reverses. The current will flow from the voltage sourceVs through Co2, Do2 and through Lr in an opposite direction to the flowin the previous cycle. The load is applied across the series combinationformed by Co1 and Co2. This circuit maintains the same advantages of thecircuit depicted in FIG. 4A. The voltage across each rectifier isclamped to the output voltage. The current through rectifiers reacheszero prior to the application of a reverse voltage across them. As aresult the reverse recovery losses are eliminated. If the converteroperates in critical conduction mode, wherein the voltage polarity of 46changes after the current through 48 reaches zero, with a delaynecessary for the voltage across the rectifier which will conduct atnext cycle reaches zero or close to zero, we can reach zero voltage ornear zero voltage commutation for the rectifiers. The circuit depictedin FIG. 5A can also operate in continuous conduction mode as the circuitdepicted in FIG. 6A. In FIG. 12A the rectifiers 134 and 136 are replacedby two controlled synchronous rectifiers 810 and 812. The controlsignals Vc1 and Vc2 are depicted in FIG. 12B in 708 and 710.

[0070] In order to produce the Vs in the secondary of a transformerthere are presented three circuits, which are suitable to produce such asource. One of these circuits is presented in FIG. 8A. The timingdiagram associated with this circuit is presented in FIG. 8B. Thiscircuit is familiar to those skilled in the art. It is known as phaseshift full bridge. It is formed by two complementary half bridges, oneformed by M1 and M2 and another one formed by M3 and M4. The controlsignals for M1 and M4 are presented in 80. The controlled signals for M3and M4 are presented in 82. During the conduction of M1 and M4 the inputvoltage is applied to the primary winding 110 of the transformer T2. Thevoltage induced in the secondary winding 112 is positive in reference tothe arrow 116. In the secondary winding 112 there will be a voltagereferred previously as 154. When the M4 turns OFF the current willcontinue to flow through the primary winding 110, and further throughthe body diode of M3, creating zero voltage switching condition for M3which is turned on at zero voltage. During the time when M1 and M3conduct, the primary winding of 106 is shorted. In the secondary, the Vswill be zero and a short-circuit characteristics.

[0071] This is equivalent to what previously was referred as 156. At themoment when M1 turns OFF the current will continue to flow through 110and the drain to source capacitance of M2 creating zero voltage or nearzero voltage conditions for M2. The voltage applied to primary winding110 will change the polarity applying a negative voltage in reference tothe arrow 112. This is equivalent to what previously was referred as158. When M3 is turned OFF the current continue to flow through 110discharging the drain to source capacitance of M4 to zero or near zero,creating zero or near zero voltage switching conditions for M4. When M2and M4 conduct the primary winding 110 is shorted and in the secondarythe state of Vs is as short circuit 156.

[0072] The circuit presented in FIG. 8A , which is know as phase shiftedfull bridge converter can generate in the secondary of the transformerthe voltage source used in describing our embodiments. Whatdifferentiate the circuit of FIG. 8A from other circuits which cangenerate a voltage source, is the short circuit behavior 156 during thetime when the voltage in secondary is zero.

[0073] The combinations of the circuits presented in FIG. 4A, 6A, 7A 5A,10A, 11A and 12A with the full bridge phase shifted topology depicted in8A, has another advantage. The slow rising the current through Lr in thesecondary will allow the full swing towards zero voltage across all theswitchers in the primary. A fast current ramp in the secondary windingwhich is specific to the prior art topologies as depicted in 2 and 3Athe soft switching in the primary is difficult to achieve for one of thecomplementary half bridge. This is due to the fact that the fast rise ofthe current in the secondary will steal some of the primary currentflowing through the resonant tank formed by the magnetizing inductanceof the transformer and the parasitic capacitance of the switchers.

[0074] Another circuit capable to provide the secondary voltage Vs withthe bydirectional low impedance characteristics during 156, is depictedin FIG. 9A. This topology is not known by those skilled in the art. Thistopology is a modification of a conventional half bridge converter withthe addition of a supplementary bydirectional auxiliary switch S1. Thetiming diagram is depicted in FIG. 9B. The switching elements 118 and120 are controlled by the signals 122 and 124. The control signals 122and 124 have the same duration, in between these two signals is a deadtime. By increasing the duration of 122 and 124 and accordinglydecreasing the duration of the dead time, the power transferred to theoutput can be controlled. An additional control signal 132 controls thebydirectional switch S1. The control signal 132 is turning the switch334 ON during the dead time 910. There is a dead time between thefalling edge of 122 and the rising edge of 132. There is also a deadtime between the falling edge of 132 and rising edge of 124. This delaytime is necessary to allow the voltage across the switching elements118, 120 and 334 to swing in order to achieve zero voltage-switchingconditions. In FIG. 9B as is depicted on 88, the voltage across theswitching element 120, VM2, has a soft transition from Vin level to avoltage plateau Vin/2 during the conduction of SI and further a softcommutation to zero after the falling edge of 132. The current throughswitching element 120 is depicted on 90. During the conduction of 118there is a voltage in the secondary winding 112 of the transformer T2.This is equivalent to 154. During the conduction of 120 there is anegative voltage across 112 , equivalent to 158. During the conductionof 132, there is a short circuit across the primary winding 110, whichreflects in the secondary winding 112. This state is equivalent to 156.

[0075] The topology described in FIG. 9A provides in the secondarywinding 112 of the transformer 106 the voltage source with thecharacteristics required in our embodiments. In addition to this, thecircuit of FIG. 9A offers zero voltage switching conditions for bothswitching elements, and recycles the leakage inductance energy which isnot dissipate, but used for discharging the parasitic capacitance of118, 120 and 334.

[0076] A third circuit capable to produce the required voltage sourcecharacteristics of Vs, is presented in FIG. 14A. There are two powertrains, formed by two transistor forward topologies. The first powertrain contains two switching elements M11 and M12, controlled by thesame control signal Vc11, 968. The first power train contains also atransformer T11, 988. The second power train contains two switchingelements M13 and M14. Both switching elements are controlled by the samecontrol signal, Vc13, 972. In the second power train there is atransformer T12, 990, which has the secondary winding 980 in series withthe secondary winding 978 of the transformer 988. The timing for Vc11and Vc13 is presented in FIG. 14B, on 992 and 994. The power ismodulating by the phase shift between the Vc11 and Vc13. The voltage inthe secondary of the transformers 988 and 990 will substract during theoverlapping time of Vc11 and Vc13, creating in the secondary the 156signal. In the secondary in series with the secondary windings 978 and980, there is the soft commutation inductive element Lr. The secondaryrectifier means and the output filter is the one described in FIG. 4A.

[0077] In FIG. 1A is depicted a circuit wherein the embodiment of claim4A is combined with the circuit described in FIG. 8A. There is anadditional circuit formed by an inductor element 440 and a capacitor442, The additional circuit creates a triangular current waveform whichis superimposed on the currents through M1 and M2. In FIG. 1B ispresented the timing diagrams of the key waveforms of the circuitillustrated in FIG. 1A. The control signals for M1 and M2 are presentedon 914. The control signal for M3 and M4 is presented on 916. Thetriangular shaped additional current 922 flowing through 440 and 442 arepresented on 918. The current 924 flowing through M1 is the result ofthe superposition of the 922 and the current reflected from thesecondary of the transformer. The presence of 918 allows zero voltageswitching conditions for M1 and M2. The additional current 922 will addto the magnetizing current of transformer T2 and discharge the parasiticcapacitance of M1 and M2 prior the switchers M1 and M2 are turned ON.The magnitude of 922 is controlled by the size of 440. A lowerinductance of 440 will increase the additional current 922 . This willensure the zero voltage switching conditions for M1 and M2. Theswitchers M3 and M4 have an inherent zero voltage switchingcharacteristics. If zero voltage switching has to be reached even atzero phase shift on both section of the full bridge, a similar circuitformed of an inductor in series with a capacitor can be inserted betweenthe GND and the M3 and M4 at the node where the transformer T2 isinserted. The voltage across the 442 and the additional capacitor is thesame and equal to Vin/2. As a result the circuit can be simplified byconnecting only one inductor with center tap across the primary winding110. The center tap of the additional inductor can be further connectedto a capacitor which has the second termination connected to the GND.The capacitor 442 can be also formed by two capacitors in series oneconnected to the positive end of the Vin and the second capacitorconnected to the negative end of 130.

[0078] The common node of these capacitors is connected to 440. In FIG.13A is presented the combination of the circuit presented in FIG. 4A andthe full bridge phase shifted circuit depicted in FIG. 8A. The circuitpresents another embodiment of the invention wherein the inductorelement Lr 48 is transferred in the primary of the transformer T2. Themode of operation is similar with the circuit wherein the inductorelement Lr is located in the secondary of the transformer. One advantageof this circuit is the fact that the current flowing through Lr willhelp to achieve zero voltage switching conditions for the primaryswitchers 92, 96, 94 and 160, with the penalty of an increase in theflux density in the transformer's core 108.

[0079] In FIG. 13B the soft commutation inductor is split in twoelements, one in the primary of the transformer 48A and one in thesecondary of the transformer 48B. The ratio between 48B and 48Areflected in the secondary can be chosen for the optimization of thecircuit. The optimization will be chosen for different criteria functionof the priority of the design. It is important to understand that Lr,Lr1 or Lr2 can be implemented by the leakage inductance of thetransformer. An additional discrete inductive element in series with theequivalent leakage inductance may or may not be necessary, function ofthe application.

[0080] In FIG. 13C the soft commutation inductor element is split in twosection 48C and 48D. These two sections are implemented on the samemagnetic core 930. An additional capacitor Czvs 932 is inserted inbetween the 48C and 48D and the ground. The same effect can be reachedif the 932 will be connected to positive end of 130.

[0081] There are two transformers T2 and T600, which have the primarywindings 110 and 606 in series and the secondary windings 620 and 608also in series. The invention does not limit to two transformers. It canbe a number of transformers, preferable an even number and theconnection to the capacitor 932 will be done in the middle having anequal number of transformers at each side of the connection.

[0082] This circuit formed by 932, 48C and 48D will add supplementarycurrents, which will assist in achieving, zero voltage switching for 92,96 , 94 and 160. When the diagonal switchers are conducting such as 94and 96 or 92 and 160, the combination LrC and LrD will exhibit a higherimpedance calculated in a such way to achieve optimum energy transferredto the secondary as presented in FIG. 13A. When the upper switchers 92and 94 or the lower switchers 96 and 104 conduct the impedance betweenthe end of 932 not connected to the GND, and the transformers T2 andT600 primary winding is very small. This will lead to circulatingcurrents through 932, which will allow zero voltage switching conditionseven at zero phase shift. This is very important in applications whereinzero voltage switching can be accomplished regardless of the phaseshift.

[0083] In FIG. 13D is depicted a circuit wherein the soft commutationinductor 48, is transferred to the secondary, and the LrC and LrD issubstituted by Lzv 440. The combination 440 and 442 is connected inbetween the T2 and T600. This circuit has the advantage of providing anadditional triangular current through both sections of the full bridge,M1 and M2 and also M3 and M4. This structure can offer zero voltageswitching conditions on all four switching elements, 92, 96, 94 and 160regardless of the phase shift, even at zero phase shift. This circuitoffer significant advantages over the prior art, such as soft switchingacross the rectifiers 50, 52, 54 and 56, and also across all theswitching elements in the primary regardless of load, input voltage andphase shift.

[0084] Many alterations and modifications may be made by those havingordinary skill in the art without departing from the spirit of theinvention. For example, is the use of several transformers on each sideof the connection between 440, 616, and 606. The capacitor 442 can beimplemented by using two capacitors in series which have the non-commonnode connected to each end of Vin, 130.

[0085] In FIG. 15 is presented a packaging concept suitable with thisinvention. All the switching elements such as 200 a, 200 b, 200 c, 200 dand 214 a, 214 b, 214 c and 214 d, are attached on a multilayers board202, and cooled by means of via or thermally conductive inserts locatedunder the switching elements to a base plate 932 attached under themultilayers boards 202. In between the base plate 932 and multilayersboard 202 there is a thermally conductive insulation material 934. Themagnetic elements are constructed using spiral traces inside of themultilayer board 202 with cutouts 218, to allow the magnetic cores 216to penetrate through and to close the magnetic circuit with secondmagnetic core 936 attached from the bottom of the 202. The thermallyconductive plate is interrupted under the magnetic core or it canprovide cavities to accommodate them. A supplementary soft elasticmaterial 938 with good thermal conductivity is inserted in between thecore and the metal plate. Some additional electronic components such as210, 208 can also be placed on 202. Pressed connectors such as 204 a,204 b, 206 b and 206 c can be inserted in 202 to offer a low impedancepath for the input and the output current to an external mother PCB. Theadvantage of this packaging is the reduction of the stray impedanceassociated with the interconnection between the switching elements andthe magnetic elements. It offers also a solid mechanical constructionsuitable for demanding working environment conditions.

[0086] The invention is defined by the following claims wherein may besubstituted therein for obtaining substantially the same result evenwhen not obtained by performing substantially the same function insubstantially the same way.

I claim:
 1. An AC-DC converter comprising of: a low impedance AC sourceproviding an alternation of a positive voltage, a negative voltage and adead time; an inductive element connected in series with said AC source;a bridge of rectifiers means having a first input terminal a secondinput terminal a first output terminal and a second output terminal; thefirst rectifier means being connected between the first input terminaland the first output terminal with the cathode to the first outputterminal, the second rectifier means being connected between the secondinput terminal and the first output terminal, with the cathode to thefirst output terminal, the third rectifier means being connected betweenthe second input terminal and the second output terminal with thecathode to the second input terminal, the fourth rectifier means beingconnected between the first input terminal and the second outputterminal with the cathode to the first input terminal, the said ACvoltage source in series with said inductive element is connectedbetween the first input terminal and the second input terminal; acapacitor in parallel with a load is connected between the first outputterminal and the second output terminal; the said voltage sourcemodulating the energy transfer through said inductor element and saidrectifiers to said capacitors and said load by changing the ratiobetween the duration of said positive and negative alternation and therepetition period of the signal provided by said voltage source.
 2. Theconverter of claim 1 wherein said the third rectifier means and said thefourth rectifier means are replaced by controlled synchronousrectifiers.
 3. The converter of claim 1 wherein all the said rectifiersmeans are replaced by controlled synchronous rectifiers.
 4. Theconverter of claim 1 wherein the current flowing through said inductiveelement reaches zero level before the voltage produced by said voltagesource changes its polarity.
 5. The converter of claim 1 wherein thecurrent flowing through said inductive element does not reach zero levelbefore the voltage produced by said voltage source changes its polarity.6. The converter of claim 1 wherein the said voltage source changes itspolarity after the current through said inductive element reaches zeroand delayed until the voltage across the rectifiers which will conducton the next cycle reaches zero voltage.
 7. The converter of claim 1wherein an additional by-directional switch is connected between saidfirst input terminal and second input terminal, turned ON and OFF by acontrol voltage synchronized with said AC voltage source and modulatingthe power transferred to said load by modulating the conduction time. 8.The circuit of claim 1 wherein said AC voltage source is generated bythe secondary winding of a transformer having primary and said secondarywinding; the said primary winding being connected to first and secondprimary output terminal; a DC voltage source; a bridge switching circuitfor producing a chopped voltage from said DC input voltage, saidswitching circuit having a said first primary input terminal, a saidprimary second input terminal, a said first primary output terminal, anda said second primary output terminal, said first and second primaryinput terminals being adapted for connection to said DC input voltage,said switching circuit including a first switching element (M1) having aparasitic drain to source capacitance C1, M1 being connected betweensaid first primary input terminal and said first primary outputterminal, a second switching element (M2) having a parasitic drain tosource capacitance C2, M2 being connected between said second primaryinput terminal and said first primary output terminal, a third switchingelement (M3) having a parasitic drain to source capacitance C3, M3 beingconnected between said first primary input terminal and said secondprimary output terminal, and a fourth switching element (M4) having aparasitic drain to source capacitance C4, M4 being connected betweensaid second primary input terminal and said second primary outputterminal; means for determining and controlling a conduction intervalfor each M1-M4 to produce a first half-cycle, including an on-timewherein M1 and M4 is conducting and a dead time wherein M1 and M3 isconducting, a second half-cycle, including an on-time wherein M3 and M2is conducting and a dead time wherein M2 and M4 is conducting; the turnon of said switchers M1-M4 is performed when the voltage across C1-C4reaches a desired level.
 9. The converter of claim 8 wherein saidinductor element is the leakage inductance of said transformer.
 10. Theconverter of claim 8 wherein said inductor element is transferred frombeing in series with said secondary winding of said transformer to beconnected in series with said primary winding of said transformer. 11.The converter of claim 8 wherein said inductor element is split into twosections, the first said section connected in series with said secondarywinding and said second section in series with said primary winding ofsaid transformer.
 12. The converter of claim 8 wherein said inductorelement is split into two sections, the first said section connected inseries with said secondary winding and said second section in serieswith said primary winding of said transformer, one of the section can bethe leakage inductance if the said transformer.
 13. The converter ofclaim 12 wherein an additional inductor element is connected to saidfirst primary output terminal and the second termination of saidadditional inductor element is connected to an additional capacitorwhich has the termination not connected to said additional inductorelement connected to said second primary input terminal; said additionalinductor element and said additional capacitor having a resonantfrequency much lower than the operation frequency of said bridgecircuit.
 14. The converter of claim 12 wherein an additional inductorelement is connected to said second primary output terminal and thesecond termination of said additional inductor element is connected toan additional capacitor which has the termination not connected to saidadditional inductor element connected to said second primary inputterminal; said additional inductor element and said additional capacitorhaving a resonant frequency much lower than the operation frequency ofsaid bridge circuit.
 15. The converter of claim 12 the transformer isformed by an even number of transformers which have the primary windingsin series and the secondary winding in series; The said inductor elementis connected in series with the primary winding, inserted in between theeven number of transformers parting two section of transformers, eachsection having an equal number of total added turns in primary and anequal number of total turns in secondary; the center tap of saidinductor element being further connected to a capacitor; the secondtermination of the capacitor being connected to said first primary inputterminal.
 16. The converter of claim 12 the transformer is formed by aneven number of transformers which have the primary windings in seriesand the secondary winding in series; The said inductor element isconnected in series with the secondary winding, an additional inductorelement is connected with one termination in between the even number oftransformers parting two section of transformers, each section having anequal number of total added turns in primary and an equal number oftotal turns in secondary; the second termination of said additionalinductor element is connected to a capacitor further connected to thesecond termination of to said primary first input terminal.
 17. Thecircuit of claim 1 wherein said AC voltage source is generated by thesecondary winding of a transformer having primary and said secondarywinding; the said primary winding having a positive and a negativetermination; a DC voltage source having a positive and a negative end;two switching elements which are controlled ON and OFF to connect saidprimary winding of said transformer to said DC voltage source; twocapacitors connected in series and across the said DC voltage source,the common node of said capacitors connected to one end of said primarywinding; a bydirectional-switching element connected across the saidprimary winding of said transformer; said two of switching elementsconnecting said termination of said primary winding not connected tosaid capacitors to positive end of said DC voltage source for a timeperiod Ton, and after a time interval Td to the negative end of said DCvoltage source for the same time period equal to Ton. said bydirectionalswitching element connecting both said termination of said primarywinding together for a time period slight shorter than Td to allow thevoltage across said two switching elements and across said bydirectionalswitch to reach zero before are turned on.
 18. The circuit of claim 1wherein said AC voltage source is generated by the secondary windings oftwo transformers, a first transformer and a second transformer, eachhaving primary and said secondary windings; said secondary windings ofsaid transformers are connected in series; the said primary winding ofthe first transformer is connected to first and second primary outputterminal; the said primary winding of the second transformer isconnected to third and fourth primary output terminal; a DC voltagesource having a positive and a negative terminal; a bridge switchingcircuit for producing a chopped voltage from said DC input voltage, saidswitching circuit having a said first primary input terminal, a saidprimary second input terminal, a said third primary input terminal, anda said fourth primary input terminal, said first, second, third andfourth primary input terminals being adapted for connection to said DCinput voltage, said switching circuit including a first switchingelement (M11) having a parasitic drain to source capacitance C11, M11being connected between positive terminal of DC voltage source and saidfirst primary input terminal, a second switching element (M12) having aparasitic drain to source capacitance C12, M12 being connected betweensaid second primary input terminal and negative terminal of DC voltagesource, a third switching element (M13) having a parasitic drain tosource capacitance C13, M13 being connected between positive terminal ofDC voltage source and said third primary input terminal, and a fourthswitching element (M14) having a parasitic drain to source capacitanceC14, M14 being connected between said fourth primary input terminal andnegative terminal of DC voltage source; means for determining andcontrolling a conduction interval for each M11-M41 to produce a firsthalf-cycle, including an on-time wherein M11 and M12 are conducting andM13 and M14 are not conducting, a dead time wherein M11, M12, M13 andM14 are conducting, a second half-cycle , including an on-time whereinM13 and M14 are conducting and M11 and M12 are not conducting ,a deadtime wherein neither switching element M11, M12, M13 and M14 is notconducting; the turn on of said switches M11-M14 is performed when thevoltage across C11-C14 reaches a desired level.
 19. An AC-DC convertercomprising of: a low impedance AC sources providing an alternation of apositive voltage, a negative voltage and a dead time; an inductiveelement connected in series with said AC source; a bridge of rectifiersmeans and capacitors having a first input terminal a second inputterminal a first output terminal and a second output terminal; the firstrectifier means being connected between the first input terminal and thefirst output terminal with the cathode to the first output terminal, thesecond rectifier means being connected between the first input terminaland the second output terminal, with the cathode to the first inputterminal, the first capacitor connected in between the first outputterminal and the second input terminal, and the second capacitorconnected between the second input terminal and the second outputterminal; The said AC voltage source in series with said inductiveelement is connected between the first input terminal and the secondinput terminal; a load is connected between the first output terminal tothe second output terminal; the said voltage source modulating theenergy transfer through said inductor element and said rectifiers tosaid capacitors and said load by changing the ratio between the durationof said positive and negative alternation and the repetition period ofthe signal provided by said voltage source.
 20. The converter of claim19 wherein said the first rectifier means and said the second rectifiermeans are replaced by controlled synchronous rectifiers.
 21. Theconverter of claim 19 wherein the current flowing through said inductiveelement reaches zero level before the voltage produced by said voltagesource changes its polarity.
 22. The converter of claim 19 wherein thecurrent flowing through said inductive element does not reach zero levelbefore the voltage produced by said voltage source changes its polarity.23. The converter of claim 19 wherein the said voltage source changesits polarity after the current through said inductive element reacheszero and delayed until the voltage across the rectifiers which willconduct on the next cycle reaches zero voltage.
 24. The converter ofclaim 19 wherein an additional by-directional switch is connectedbetween said first input terminal and second input terminal, turned ONand OFF by a control voltage synchronized with said AC voltage sourceand modulating the power transferred to said load by modulating theconduction time.
 25. The circuit of claim 19 wherein said AC voltagesource is generated by the secondary winding of a transformer havingprimary and said secondary winding; the said primary winding beingconnected to first and second primary output terminal; a DC voltagesource; a bridge switching circuit for producing a chopped voltage fromsaid DC input voltage, said switching circuit having a said firstprimary input terminal, a said primary second input terminal, a saidfirst primary output terminal, and a said second primary outputterminal, said first and second primary input terminals being adaptedfor connection to said DC input voltage, said switching circuitincluding a first switching element (M1) having a parasitic drain tosource capacitance C1, M1 being connected between said first primaryinput terminal and said first primary output terminal, a secondswitching element (M2) having a parasitic drain to source capacitanceC2, M2 being connected between said second primary input terminal andsaid first primary output terminal, a third switching element (M3)having a parasitic drain to source capacitance C3, M3 being connectedbetween said first primary input terminal and said second primary outputterminal, and a fourth switching element (M4) having a parasitic drainto source capacitance C4, M4 being connected between said second primaryinput terminal and said second primary output terminal; means fordetermining and controlling a conduction interval for each M1-M4 toproduce a first half-cycle, including an on-time wherein M1 and M4 isconducting and a dead time wherein M1 and M3 is conducting, a secondhalf-cycle, including an on-time wherein M3 and M2 is conducting and adead time wherein M2 and M4 is conducting; the turn on of said switchesM1-M4 is performed when the voltage across C1-C4 reaches a desiredlevel.
 26. The converter of claim 25 wherein said inductor element isthe leakage inductance of said transformer.
 27. The converter of claim25 wherein said inductor element is transferred from being in serieswith said secondary winding of said transformer to be connected inseries with said primary winding of said transformer.
 28. The converterof claim 25 wherein said inductor element is split into two sections,the first said section connected in series with said secondary windingand said second section in series with said primary winding of saidtransformer.
 29. The converter of claim 25 wherein said inductor elementis split into two sections, the first said section connected in serieswith said secondary winding and said second section in series with saidprimary winding of said transformer, one of the section can be theleakage inductance if the said transformer.
 30. The converter of claim25 wherein an additional inductor element is connected to said firstprimary output terminal and the second termination of said additionalinductor element is connected to an additional capacitor which has thetermination not connected to said additional inductor element connectedto said second primary input terminal; said additional inductor elementand said additional capacitor having a resonant frequency much lowerthan the operation frequency of said bridge circuit.
 31. The converterof claim 25 wherein an additional inductor element is connected to saidsecond primary output terminal and the second termination of saidadditional inductor element is connected to an additional capacitorwhich has the termination not connected to said additional inductorelement connected to said second primary input terminal; said additionalinductor element and said additional capacitor having a resonantfrequency much lower than the operation frequency of said bridgecircuit.
 32. The converter of claim 25 the transformer is formed by aneven number of transformers which have the primary windings in seriesand the secondary winding in series; The said inductor element isconnected in series with the primary winding, inserted in between theeven number of transformers parting two section of transformers, eachsection having an equal number of total added turns in primary and anequal number of total turns in secondary; the center tap of saidinductor element being further connected to a capacitor; the secondtermination of the capacitor being connected to said first primary inputterminal.
 33. The converter of claim 25 the transformer is formed by aneven number of transformers which have the primary windings in seriesand the secondary winding in series; The said inductor element isconnected in series with the secondary winding, an additional inductorelement is connected with one termination in between the even number oftransformers parting two section of transformers, each section having anequal number of total added turns in primary and an equal number oftotal turns in secondary; the second termination of said additionalinductor element is connected to a capacitor further connected to thesecond termination of to said primary first input terminal.
 34. Thecircuit of claim 25 wherein said AC voltage source is generated by thesecondary winding of a transformer having primary and said secondarywinding; the said primary winding having a positive and a negativetermination; a DC voltage source having a positive and a negative end;two switching elements which are controlled ON and OFF to connect saidprimary winding of said transformer to said DC voltage source; twocapacitors connected in series and across the said DC voltage source,the common node of said capacitors connected to one end of said primarywinding; a bydirectional-switching element connected across the saidprimary winding of said transformer; said two of switching elementsconnecting said termination of said primary winding not connected tosaid capacitors to positive end of said DC voltage source for a timeperiod Ton, and after a time interval Td to the negative end of said DCvoltage source for the same time period equal to Ton. said bydirectionalswitching element connecting both said termination of said primarywinding together for a time period slight shorter than Td to allow thevoltage across said two switching elements and across said bydirectionalswitch to reach zero before are turned on.
 35. The circuit of claimwherein said AC voltage source is generated by the secondary windings oftwo transformers, a first transformer and a second transformer, eachhaving primary and said secondary windings; said secondary windings ofsaid transformers are connected in series; the said primary winding ofthe first transformer is connected to first and second primary outputterminal; the said primary winding of the second transformer isconnected to third and fourth primary output terminal; a DC voltagesource having a positive and a negative terminal; a bridge switchingcircuit for producing a chopped voltage from said DC input voltage, saidswitching circuit having a said first primary input terminal, a saidprimary second input terminal, a said third primary input terminal, anda said fourth primary input terminal, said first, second, third andfourth primary input terminals being adapted for connection to said DCinput voltage, said switching circuit including a first switchingelement (M11) having a parasitic drain to source capacitance C11, M11being connected between positive terminal of DC voltage source and saidfirst primary input terminal, a second switching element (M12) having aparasitic drain to source capacitance C12, M12 being connected betweensaid second primary input terminal and negative terminal of DC voltagesource, a third switching element (M13) having a parasitic drain tosource capacitance C13, M13 being connected between positive terminal ofDC voltage source and said third primary input terminal, and a fourthswitching element (M14) having a parasitic drain to source capacitanceC14, M14 being connected between said fourth primary input terminal andnegative terminal of DC voltage source; means for determining andcontrolling a conduction interval for each M11-M41 to produce a firsthalf-cycle, including an on-time wherein M11 and M12 are conducting andM13 and M14 are not conducting, a dead time wherein M11, M12, M13 andM14 are conducting, a second half-cycle, including an on-time whereinM13 and M14 are conducting and M11 and M12 are not conducting, a deadtime wherein neither switching element M11, M12, M13 and M14 is notconducting; the turn on of said switches M11-M14 is performed when thevoltage across C11-C14 reaches a desired level.